Radar device

ABSTRACT

A radar device employs a configuration provided with: a receiver which, in operation, receives one or more radar transmission signals transmitted from another radar device, in an interference measurement segment in which transmission of a radar transmission signal from the radar device is stopped; A/D conversion circuitry which, in operation, converts the one or more radar transmission signals from the other radar device received by the receiver from one or more analog signals into one or more digital signals; and an interference detection circuitry which, in operation, performs a correlation calculation between each of one or more discrete samples that is the one or more digital signals and a prescribed coefficient sequence to detect one or more prescribed frequency components included in the one or more digital signals, as one or more interference signal components.

BACKGROUND

1. Technical Field

The present disclosure relates to a radar device that detectsinterference.

2. Description of the Related Art

In recent years, high-resolution radar that uses microwaves, milliwaves,and the like has been studied. Furthermore, the development ofwide-angle radar that detects not only vehicles but also pedestrians isneeded to improve safety outdoors.

In wide-angle pulse radar that detects vehicles and pedestrians, aplurality of reflected waves from near-distance targets (for example,vehicles) and far-distance targets (for example, people) are mixedwithin reception signals, and therefore radar transmission units arerequired to have a transmission configuration that transmits pulse wavesor pulse modulated waves having low range sidelobe characteristics.Furthermore, radar reception units are required to have a receptionconfiguration that has a broad reception dynamic range.

Proposals have been made for pulse compression radar that uses Barkercodes, M-sequence codes, and complementary codes as pulse waves or pulsemodulated waves for obtaining low range sidelobe characteristics. Inparticular, a method for generating complementary codes is disclosed in‘New Complementary Pairs of Sequences’, Budisin S. Z., Electron. Lett.,1990, 26, (13), pp. 881-883.

Complementary codes can be generated as follows, for example. To bespecific, complementary codes having a code length of L=4, 8, 16, 32, .. . , 2^(P) can be sequentially generated on the basis of code sequencesof a=[1 1] and b=[1 −1] that are complementary and are made up of theelements 1 or −1. Although the required reception dynamic rangeincreases as the code length increases, with complementary codes, thepeak sidelobe ratio (PSR) can be reduced with a shorter code length.Therefore, the dynamic range required for reception can be reduced evenin the case where a plurality of reflected waves from near-distancetargets and far-distance targets are mixed. However, in the case whereM-sequence codes are used, the PSR is given at 20 log(1/L), and in orderto obtain a low range sidelobe, a code length L that is longer than thatof a complementary code becomes necessary (for example, L=1024 in thecase where the PSR=60 dB).

In the case where the frequency bands of radio waves output by aplurality of radar devices are the same band or some of the bandsoverlap, interference among the radar devices occurs when a positionalrelationship develops in which the detection areas of the plurality ofradar devices overlap. In other words, a relationship develops in whichthe radio waves output by a certain radar device are received by anotherradar device. Interference between the radar devices become stronginterference as the positional relationship between the radar devicesbecomes closer (in other words, as the distance therebetween decreases),the non-detection rate or the erroneous detection rate increases fortargets that should originally be detected, and deterioration indetection performance increases.

Therefore, a technique that prevents deterioration in detectionperformance caused by interference between radar devices, by detectinginterference components from another radar device is disclosed inJapanese Unexamined Patent Application Publication No. 2006-220624, forexample.

Japanese Unexamined Patent Application Publication No. 2006-220624discloses a device that determines interference from another radardevice mounted in a vehicle. With vehicle-mounted radar, the detectionarea changes as the vehicle travels. In the case where the frequencybands of output radio waves are the same or some of the bands overlapbetween vehicle-mounted radar devices mounted in a plurality ofvehicles, interference occurs when a positional relationship develops inwhich the detection areas overlap.

With regard to this kind of interference, Japanese Unexamined PatentApplication Publication No. 2006-220624 discloses a configuration thatis a reception configuration for a frequency modulated continuous wave(hereinafter referred to as FMCW) radar device and detects interferencefrom another FMCW radar device. An FMCW radar device uses frequencyspectrum data of obtained beat signals to obtain an integral strengthvalue in a prescribed frequency range, and determines that interferencewith another radar device has occurred in the case where the integralstrength value exceeds an interference determination threshold value.

SUMMARY

In the aforementioned FMCW radar device disclosed in Japanese UnexaminedPatent Application Publication No. 2006-220624, reflected waves of radiowaves output by the radar device are also included in the calculatedstrength integral value, and the amount thereof depends upon thesituation such as the surrounding structures or the road surface.Therefore, in order to suppress erroneous interference determinations,it is necessary to set a determination threshold value to besufficiently high, and there is a possibility of there being a decreasein interference detection sensitivity.

One non-limiting and exemplary embodiment provides a radar device thatimproves detection sensitivity for interference from another radardevice.

In one general aspect, the techniques disclosed here feature: a radardevice provided with: a receiver which, in operation, receives one ormore radar transmission signals transmitted from another radar device,in an interference measurement segment in which transmission of one ormore radar transmission signals from the radar device is stopped; an A/Dconverting circuitry which, in operation, converts the one or more radartransmission signals from the other radar device received by thereceiver from one or more analog signals into one or more digitalsignals; and an interference detecting circuitry which, in operation,performs a correlation calculation between each of the one or morediscrete samples that are the one or more digital signals and aprescribed coefficient sequence to detect one or more prescribedfrequency components included in the one or more digital signals, as oneor more interference signal components.

It should be noted that general or specific embodiments may beimplemented as a system, a method, an integrated circuit, a computerprogram, a storage medium, or any selective combination thereof.

According to the present disclosure, it is possible to improve detectionsensitivity for interference from another radar device.

Additional benefits and advantages of the disclosed embodiments willbecome apparent from the specification and drawings. The benefits and/oradvantages may be individually obtained by the various embodiments andfeatures of the specification and drawings, which need not all beprovided in order to obtain one or more of such benefits and/oradvantages.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram depicting the configuration of a radar deviceaccording to embodiment 1 of the present disclosure;

FIG. 2 is a drawing depicting the way in which switching is performedbetween an interference measurement segment and a distance measurementsegment;

FIG. 3 includes a drawing depicting radar transmission signals ofdistance measurement segments, and a drawing depicting radartransmission signals of interference measurement segments;

FIG. 4 is a block diagram depicting the internal configuration of aninterference detection unit of FIG. 1;

FIG. 5 is a block diagram depicting the internal configuration of afrequency component extraction unit of FIG. 4;

FIG. 6 is a drawing depicting transmission timings of a radartransmission signal and reception timings of a reflected wave;

FIG. 7 is a drawing depicting a relationship between the arrangement ofreception antenna elements that make up an array antenna and an azimuthangle;

FIG. 8 is a drawing depicting a relationship between a radar signal bandand an interference wave-detection frequency component;

FIG. 9 is a drawing depicting an FMCW modulated wave of another radardevice;

FIG. 10 is a drawing depicting the output of an interference detectionunit;

FIG. 11 is a block diagram depicting the internal configuration of aninterference detection unit according to embodiment 2 of the presentdisclosure;

FIG. 12 is a block diagram depicting the configuration of a radar deviceaccording to embodiment 3 of the present disclosure;

FIG. 13 is a block diagram depicting the configuration of a radar deviceaccording to modified example 1 of the present disclosure; and

FIG. 14 is a block diagram depicting the internal configuration of aradar transmission signal generation unit according to modified example2 of the present disclosure.

DETAILED DESCRIPTION

Hereafter, embodiments of the present disclosure will be described indetail with reference to the drawings. However, in the embodiments,configurations having the same function are denoted by the samereference numbers and redundant descriptions are omitted.

Embodiment 1

FIG. 1 is a block diagram depicting the configuration of a radar device10 according to embodiment 1 of the present disclosure. The radar device10 is provided with a radar transmission unit 20, a radar reception unit30, a reference signal generation unit 11, a transmission control unit12, and an interference countermeasure control unit 13.

First, the configuration of the radar transmission unit 20 will bedescribed.

The radar transmission unit 20 is provided with a radar transmissionsignal generation unit 21, a transmission RF unit 25, and a transmissionantenna 26. The radar transmission signal generation unit 21 is providedwith a code generation unit 22, a modulation unit 23, and a band controlfilter (denoted as “LPF” (low pass filter) in the drawing andhereinafter referred to as “LPF”) 24. Furthermore, the radartransmission signal generation unit 21 generates a timing clock producedby multiplying a reference signal from the reference signal generationunit 11 by a prescribed number, and on the basis thereof, repeatedlyoutputs a baseband radar transmission signal r(n, M)=I(n, M)+jQ(n, M) ina prescribed radar transmission period Tr. It should be noted that jrepresents an imaginary unit, n represents a discrete timepoint, and Mrepresents an ordinal number for a radar transmission period.

The code generation unit 22 generates codes a_(n) that constitute a codesequence (an M-sequence code, a Barker code sequence, a complementarycode sequence, or the like) of the code length L, and outputs to themodulation unit 23. It should be noted that n=1, . . . , L. The codesa_(n) are generated in each radar transmission period Tr.

In the case where the code sequence is a complementary code sequence(including a Golay code sequence, a Spano code sequence, or the like),codes P_(n) and Q_(n) that constitute a pair are each generatedalternately in each radar transmission period. In other words, a codeP_(n) is transmitted as a pulse compression code a_(n) in an M^(th)radar transmission period Tr, and then a code Q_(n) is transmitted as apulse compression code b_(n) in an M+1^(th) radar transmission periodTr. In the radar transmission periods thereafter (m+2^(th) . . . ),transmission is repeatedly performed in the same way with M^(th) toM+1^(th) radar transmissions serving as single units.

A complementary code is made up of two code sequences (hereinafter takenas pulse compression codes a_(n) and b_(n); furthermore, n=1, . . . , L,and L is the code sequence length). Autocorrelation calculations foreach of the pulse compression codes a_(n) and b_(n) are given in thefollowing expressions (1) and (2). When the results thereof are addedwith the shift times τ thereof being consistent (see the followingexpression (3)), a correlation value is reached with which the rangesidelobe is 0. Complementary codes have the aforementioned properties.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack & \; \\{{R_{aa}(\tau)} = {\sum\limits_{n = 1}^{L}\; {a_{n}a_{n + \tau}^{*}}}} & (1) \\\left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack & \; \\{{{R_{bb}(\tau)} = {\sum\limits_{n = 1}^{L}\; {b_{n}b_{n + \tau}^{*}}}}{{Here},{a_{n} = 0}}{and}{b_{n} = {{0\mspace{14mu} {in}\mspace{14mu} n} > {L\mspace{14mu} {and}\mspace{14mu} n} < 1.}}} & (2) \\\left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack & \; \\\left\{ \begin{matrix}{{{{R_{aa}(\tau)} + {R_{bb}(\tau)}} \neq 0},{{{when}\mspace{14mu} \tau} = 0}} \\{{{{R_{aa}(\tau)} + {R_{bb}(\tau)}} = 0},{{{when}\mspace{14mu} \tau} \neq 0}}\end{matrix} \right. & (3)\end{matrix}$

The modulation unit 23 performs pulse modulation (amplitude modulation,ASK, pulse shift keying) or phase modulation (PSK) with respect to thecode sequence output from the code generation unit 22, and outputs tothe LPF 24.

The LPF 24 outputs the modulated signal output from the modulation unit23, to the transmission RF unit 25 as a radar transmission signal of abaseband limited to within a prescribed band.

The transmission RF unit 25 converts the baseband radar transmissionsignal output from the radar transmission signal generation unit 21 intoa carrier frequency (radio frequency: RF) band by frequency conversion.Furthermore, the transmission RF unit 25 amplifies the carrierfrequency-band radar transmission signal to a prescribed transmissionpower P [dB] with a transmission amplifier and outputs to thetransmission antenna 26.

The transmission antenna 26 radiates the radar transmission signaloutput from the transmission RF unit 25 into a space.

The transmission control unit 12 performs transmission control thatdiffers in accordance with two operation segments depicted in FIG. 2, inother words, an interference measurement segment in which a radartransmission signal transmitted from another radar device is measured,and a distance measurement segment in which the distance to a target ismeasured.

FIG. 3 includes a drawing depicting radar transmission signals ofdistance measurement segments, and a drawing depicting radartransmission signals of interference measurement segments. FIG. 3(a)depicts radar transmission signals of distance measurement segments. Aradar transmission signal is present in a code transmission segment Twof each radar transmission period Tr, and the segments (Tr−Tw) thatremain are non-signal segments. Furthermore, a pulse code sequence ofthe pulse code length L is included within the code transmissionsegments Tw; however, by carrying out modulation that uses a No samplefor each single pulse code, an Nr=No×L sample signal is included withineach code transmission segment Tw. Furthermore, an Nu sample is includedin the non-signal segments (Tr−Tw) in the radar transmission periods. Onthe other hand, FIG. 3(b) depicts radar transmission signals ofinterference measurement segments. As depicted in FIG. 3(b), in theinterference measurement segments, the transmission of radartransmission signals from the radar device 10 is stopped and a state inwhich codes are not transmitted is entered for a prescribed number ofradar transmission periods.

Furthermore, the transmission control unit 12 performs transmissioncontrol in which interference measurement segments serve as N_(IM)number of code transmission periods, distance measurement segments serveas N_(RM) number of code transmission periods, and switching isperformed therebetween.

Next, the configuration of the radar reception unit 30 will bedescribed.

The radar reception unit 30 is mainly provided with antenna systemprocessing units 30 a to 30 d that correspond to the number of receptionantennas that make up an array antenna, and a direction estimation unit43. The antenna system processing units 30 a to 30 d are each providedwith a reception antenna 31, a reception RF unit 32, and a signalprocessing unit 36.

The reception antenna 31 receives a signal produced by a radartransmission signal transmitted from the radar transmission unit 20being reflected by a reflecting object including the target. A radarreception signal received by the reception antenna 31 is output to thereception RF unit 32.

The reception RF unit 32 is provided with an amplifier 33, a frequencyconversion unit 34, and a quadrature detection unit 35.

The amplifier 33 performs signal amplification with respect to the radarreception signal received by the reception antenna 31, and outputs tothe frequency conversion unit 34.

The frequency conversion unit 34 converts the radio-frequency radarreception signal output from the amplifier 33 into a low-frequency radarreception signal, and outputs to the quadrature detection unit 35.

The quadrature detection unit 35 performs quadrature detection withrespect to the low-frequency radar reception signal output from thefrequency conversion unit 34, and performs conversion into basebandsignals made up of an I signal and a Q signal. The I signal is output toan A/D conversion unit 37 a of the signal processing unit 36, and the Qsignal is output to an A/D conversion unit 37 b of the signal processingunit 36. It should be noted that a timing clock signal of the signalprocessing unit 36 for the baseband signals is generated as a timingclock of a prescribed multiple using a reference signal from thereference signal generation unit 11 in the same way as with the radartransmission signal generation unit 21.

The signal processing unit 36 is provided with the A/D conversion units37 a and 37 b, a correlation calculation unit 40, an integration unit41, a Doppler frequency analysis unit 42, an interference detection unit38, and an interference determination unit 39.

The A/D conversion units 37 a and 37 b perform sampling at discretetimes with respect to the baseband signals made up of the I signals andthe Q signals output from the quadrature detection unit 35, and performconversion into digital data. The A/D conversion units 37 a and 37 boutput the converted digital data to the correlation calculation unit 40and the interference detection unit 38. Here, for the sampling rate ofthe A/D conversion units 37 a and 37 b, Ns number of discrete samplesare performed at each one pulse time Tp (=Tw/L) in the radartransmission signal, in other words, Ns number of over-samples per onepulse. It should be noted that, hereinafter, baseband signals Ir(k, M)and Qr(k, M) made up of I signals and Q signals of a discrete timepointk in an M^(th) radar transmission period are indicated using a complexnumber x(k, M)=Ir(k, M)+jQr(k, M). Furthermore, j is an imaginary unit.Moreover, hereinafter, with regard to the timepoints k, measurement upto k=(Nr+Nu)Ns/No, which is a sample point up to prior to a radartransmission period Tr ending, is periodically performed with the timingat which the radar transmission period Tr starts serving as a reference(k=1). In other words, k=1, (Nr+Nu)Ns/No.

The interference detection unit 38 detects one or more interferencesignal components in an interference measurement segment on the basis ofa control signal from the transmission control unit 12, and outputs thedetected one or more interference signal components to the interferencedetermination unit 39. FIG. 4 is a block diagram depicting the internalconfiguration of the interference detection unit 38 of FIG. 1. In FIG.4, from the digital data output from the A/D conversion units 37 a and37 b, a frequency component extraction unit 51 extracts one or moreinterference signal components in a specific frequency componentincluded within the baseband band of the radar signals used by the radardevice 10, and outputs to a square calculation unit 52.

The square calculation unit 52 squares the one or more interferencesignal components output from the frequency component extraction unit51, and outputs to the interference determination unit 39.

The frequency component extraction unit 51, in order to extract aspecific frequency component included within the baseband band of theradar signals used by the radar device 10, performs a correlationcalculation between a discrete sample x(k, M), which is the digital dataoutput from the A/D conversion units 37 a and 37 b, and a coefficientsequence FS_(n) for extracting the specific frequency component (seeexpression (4)). Here, L_FS is the sequence length of the coefficientsequence FS_(n).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack & \; \\{{{IC}\left( {k,M} \right)} = {\sum\limits_{n = 1}^{L\_ {FS}}\; {{x\left( {{k + n - 1},M} \right)} \times {FS}_{n}}}} & (4)\end{matrix}$

By using the coefficient sequence indicated in expression (5) as thecoefficient sequence FS_(n), a ¼^(th) positive frequency component isextracted from a sampling frequency Ns/Tp of the A/D conversion units 37a and 37 b, and thus a specific frequency component Ns/(4Tp) can beextracted.

[Equation 5]

{FS ₁ ,FS ₂ ,FS ₃ ,FS ₄}={1,j,−1,−j}  (5)

Furthermore, the frequency component extraction unit 51, which uses thecoefficient sequence FS_(n) indicated in expression (5), can be realizedwith the configuration depicted in FIG. 5. The frequency componentextraction unit 51 depicted in FIG. 5 is provided with delayers 61 a to61 c, coefficient multipliers 62 a to 62 d, and an adder 63.

The delayers 61 a to 61 c delay input data and output delayed data. Thedelayer 61 a delays the complex number made up of the I signal and the Qsignal output from the A/D conversion units 37 a and 37 b, and outputs adelayed discrete sample to the coefficient multiplier 62 b and thedelayer 61 b. The delayer 61 b delays the output from the delayer 61 a,and outputs delayed data to the coefficient multiplier 62 c and thedelayer 61 c. The delayer 61 c delays the output from the delayer 61 b,and outputs a digit of delayed data to the coefficient multiplier 62 d.

The coefficient multiplier 62 a multiplies the discrete sample outputfrom the A/D conversion units 37 a and 37 b by a coefficient 1, andoutputs the multiplication result to the adder 63. The coefficientmultiplier 62 b multiplies the data output from the delayer 61 a by acoefficient j, and outputs the multiplication result to the adder 63.The coefficient multiplier 62 c multiplies the data output from thedelayer 61 b by a coefficient −1, and outputs the multiplication resultto the adder 63. The coefficient multiplier 62 d multiplies the dataoutput from the delayer 61 c by a coefficient −j, and outputs themultiplication result to the adder 63. In should be noted that j is animaginary unit.

The adder 63 adds the multiplication results output from the coefficientmultiplier 62 a to 62 d, and outputs the addition result to the squarecalculation unit 52.

Furthermore, by using the coefficient sequence indicated in expression(6) as the coefficient sequence FS_(n), a ¼^(th) negative frequencycomponent is extracted from the sampling frequency Ns/Tp of the A/Dconversion units 37 a and 37 b, and therefore a specific frequencycomponent −Ns/(4Tp) can be extracted.

[Equation 6]

{FS ₁ ,FS ₂ ,FS ₃ ,FS ₄}={1,−j,−1,j}  (6)

Furthermore, by using the coefficient sequence indicated in expression(7) as the coefficient sequence FS_(n), a ⅛^(th) positive frequencycomponent is extracted from the sampling frequency Ns/Tp of the A/Dconversion units 37 a and 37 b, and therefore a specific frequencycomponent Ns/(8Tp) can be extracted.

[Equation 7]

{FS ₁ ,FS ₂ ,FS ₃ ,FS ₄ ,FS ₅ ,FS ₆ ,FS ₇ ,FS₈}={1,exp(jπ/4),j,exp(j3π/4),−1,exp(−jπ/4),−j,exp(−j3π/4)}   (7)

Furthermore, by using the coefficient sequence indicated in expression(8) as the coefficient sequence FS_(n), a ⅛^(th) negative frequencycomponent is extracted from the sampling frequency Ns/Tp of the A/Dconversion units 37 a and 37 b, and therefore a specific frequencycomponent −Ns/(8Tp) can be extracted.

[Equation 8]

{FS ₁ ,FS ₂ ,FS ₃ ,FS ₄ ,FS ₅ ,FS ₆ ,FS ₇ ,FS₈}={1,exp(−jπ/4),−j,exp(−j3π/4),−1,exp(jπ/4),j,exp(j3π/4)}   (8)

Furthermore, by using the coefficient sequence indicated in expression(9) as the coefficient sequence FS_(n), a 1/(2G)^(th) positive frequencycomponent is extracted from the sampling frequency Ns/Tp of the A/Dconversion units 37 a and 37 b, and therefore a specific frequencycomponent Ns/(2G×Tp) can be extracted. Here, n=1, . . . , 2G.

[Equation 9]

FS _(n)=exp[jπ(n−1)/G]  (9)

Furthermore, by using the coefficient sequence indicated in expression(10) as the coefficient sequence FS_(n), a 1/(2G)^(th) negativefrequency component is extracted from the sampling frequency Ns/Tp ofthe A/D conversion units 37 a and 37 b, and therefore a specificfrequency component −Ns/(2G×Tp) can be extracted. Here, n=1, . . . , 2G.

[Equation 10]

FS _(n)=exp[−jπ(n−1)/G]  (10)

It should be noted that detection sensitivity can be improved byadditionally repeatedly using any of the aforementioned coefficientsequences. In other words, in the case where the coefficient length ofthe coefficient sequence FS_(n) for extracting a specific frequencycomponent is taken as L_FS, the detection sensitivity for the specificfrequency component can be increased N times when that coefficientsequence is repeated N times (an SNR improvement of 10 log₁₀(N) [dB]).For example, by using a coefficient sequence {1, −j, −1, j, 1, −j, −1,j} in which {FS₁, FS₂, FS₃, FS₄}={1, −j, −1, j} is repeated twice, thedetection sensitivity for the specific frequency component −Ns/(4Tp) canbe doubled.

The interference determination unit 39, on the basis of the controlsignal output from the transmission control unit 12, determines whetheror not the one or more interference signal components output from theinterference detection unit 38 in an interference measurement segmentexceeds a prescribed determination level. The interference determinationunit 39 determines that an interference component is not present in thecase where each of the one or more interference signal components isequal to or less than the determination level, and determines that aninterference component is present in the case where any of the one ormore interference signal components exceeds the determination level.

It should be noted that the interference detection unit 38 and theinterference determination unit 39 are provided in at least one antennasystem processing unit from among a first antenna system processing unitto an Na^(th) antenna system processing unit.

The interference countermeasure control unit 13 performs interferencecountermeasure control in the subsequent distance measurement segment onthe basis of the interference determination result output from theinterference determination unit 39 in the interference measurementsegment. In other words, in the case where the interferencedetermination unit 39 has determined that an interference component ispresent, in order to reduce or suppress the one or more interferencesignal components, control that uses any of the following or acombination thereof is applied in the subsequent distance measurementsegment to perform radar transmission/reception operations in thedistance measurement segment.

(1) The interference countermeasure control unit 13 performs controlthat changes the carrier frequency of the radar device 10. In otherwords, the transmission carrier frequency of the transmission RF unit 25is changed. Furthermore, it is made possible for the transmissioncarrier frequency changed by the transmission RF unit 25 to be receivedalso by the reception RF unit 32. The frequency is changed by performingcontrol that shifts a preset frequency interval. It should be notedthat, in the case where a configuration that detects a positive/negativespecific frequency component is used as the interference detection unit38, it becomes possible for the frequency signal component to be reducedor suppressed to a greater extent by changing the transmission carrierfrequency in a frequency direction in which the detectedpositive/negative frequency components are low in number. Furthermore,control may be performed that, as the detected one or more interferencesignal components increase in number, widens the frequency interval thatis used when the frequency is changed. It thereby becomes possible forthe one or more interference signal components to be reduced orsuppressed more effectively.

(2) In the case where the vertical beam direction of the transmissionantenna 26 or the reception antenna 31 of the radar device 10 can becontrolled, the interference countermeasure control unit 13 performscontrol that changes the beam direction to a downward direction for aprescribed time interval.

(3) The interference countermeasure control unit 13 performs controlthat, for a prescribed time interval, increases the code length of theradar transmission signals used by the radar device 10.

The correlation calculation unit 40, in the distance measurement segmentfollowing interference detection and interference countermeasure controlin an interference measurement segment, performs a correlationcalculation between a discrete sample x(k, M) output from the A/Dconversion units 37 a and 37 b at each radar transmission period and apulse compression code a_(n) of the code length L that is transmitted.Here, n=1, . . . , L. A sliding correlation calculation in an M^(th)radar transmission period is performed on the basis of the followingexpression (11), for example.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack & \; \\{{{AC}\left( {k,M} \right)} = {\sum\limits_{n = 1}^{L}\; {{x\left( {{k + {{Ns}\left( {n - 1} \right)}},M} \right)}a_{n}^{*}}}} & (11)\end{matrix}$

In expression (11), AC(k, M) indicates a correlation calculation valueof a discrete timepoint k. The asterisk (*) represents a complexconjugate operator. Furthermore, the calculation of AC(k, M) isperformed for a period of k=1, . . . , (Nr+Nu)Ns/No.

It is possible for the calculation in the correlation calculation unit40 to be performed for k=1, . . . , (Nr+Nu)Ns/No; however, it should benoted that the measurement range (range of k) may be additionallylimited by the presence range for the target to be measured of the radardevice 10. It thereby becomes possible for the calculation processingamount to be reduced. For example, the measurement range may be limitedto k=Ns(L+1), . . . , (Nr+Nu)Ns/No−NsL. In this case, as depicted inFIG. 6, measurement is not performed in a time segment that correspondsto a code transmission segment, and even in a case such as when a radartransmission signal directly enters the radar reception unit 30, itbecomes possible to perform measurement with the effect thereof havingbeen eliminated. In the case where the measurement range (range of k) islimited, the following processing also similarly applies processing inwhich the measurement range (range of k) is limited.

On the basis of the correlation calculation value AC(k, M), which is anoutput of the correlation calculation unit 40 for each discretetimepoint k, the integration unit 41 performs Np number of summationsfor a period (Tr×Np), which is a plural Np number of the radartransmission periods Tr, in accordance with the following expression(12).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 12} \right\rbrack & \; \\{{{CI}\left( {k,m} \right)} = {\sum\limits_{g = 1}^{Np}\; {{AC}\left( {k,{{{Np}\left( {m - 1} \right)} + g}} \right)}}} & (12)\end{matrix}$

In expression (12), Np is an integer value that is equal to or greaterthan 1. In other words, the integration unit 41 performs summation Npplurality of times with single units being constituted by the output ofthe correlation calculation unit 40 obtained with the radar transmissionperiods Tr serving as units. In other words, a correlation value CI(k,m) that is added with the timings of the discrete timepoints k beingaligned is calculated at each discrete timepoint k with AC(k, Np(m−1)+1)to AC(k, Np×m) serving as units. It should be noted that m is a naturalnumber. Thus, due to the effect of the addition, the SNR can beincreased and the measurement performance relating to estimating thearrival distance of the target can be improved, in a range in whichreception signals of reflected waves from the target have a highcorrelation, in a time range in which addition is performed Np times.

A condition with which the phase components are within a certain rangefor a segment in which addition is performed is required in order for anideal addition gain to be obtained, and the number of times thataddition is to be applied is set on the basis of an assumed maximummovement speed of the target to be measured. This is because, as theassumed maximum speed of the target increases, the time period in whichtime correlation is high becomes shorter due to the influence of Dopplerfrequency fluctuations included in the reflected waves from the target,the Np becomes a small value, and the gain improvement effect broughtabout by addition decreases.

The Doppler frequency analysis unit 42 performs coherent integrationwith CI(k, Nc(w−1)+1) to CI(k, Nc×w), which are Nc number of outputs ofthe integration unit 41 obtained at each discrete timepoint k, servingas single units, the timings of the discrete timepoints k being aligned,and a phase fluctuation Φ(fs)=2πfs(Tr×Np)ΔΦ, which corresponds to 2Nfnumber of different Doppler frequencies fsΔΦ, being corrected inaccordance with the following expression (13).

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack} & \; \\{{{FT\_ CI}^{Nant}\left( {k,{fs},w} \right)} = {{\sum\limits_{q = 0}^{{Nc} - 1}\; {{{CI}\left( {k,{{{Nc}\left( {w - 1} \right)} + q + 1}} \right)}{\exp \left\lbrack {{- j}\; {\varphi ({fs})}q} \right\rbrack}}} = {\sum\limits_{q = 0}^{{Nc} - 1}\; {{{CI}\left( {k,{{{Nc}\left( {w - 1} \right)} + q + 1}} \right)}{\exp \left\lbrack {{- j}\; 2\; \pi \; {fsTrNpq}\; \Delta \; \varphi} \right\rbrack}}}}} & (13)\end{matrix}$

In expression (13), FT_CI^(Nant)(k, fs, w) is the w^(th) output by theDoppler frequency analysis unit 42, and indicates a coherent integrationresult of the Doppler frequencies fsΔΦ at the discrete timepoints k, inthe Nant^(th) antenna system processing unit. It should be noted thatNant=1 to Na, fs=−Nf+1, . . . , 0, . . . , Nf, k=1, . . . ,(Nr+Nu)Ns/No, w is a natural number, and ΔΦ is a phase rotation unit.Thus, in each antenna system processing unit, FT_CI^(Nant)(k, −Nf+1, w),. . . , FT_CI^(Nant)(k, Nf−1, w), which are coherent integration resultsthat correspond to 2Nf number of Doppler frequency components of eachdiscrete timepoint k, are obtained for each period (Tr×Np×Nc), which isa plural Np×Nc number of the radar transmission periods Tr.

The aforementioned processing equates to the output of the integrationunit 41 being subjected to discrete Fourier transform processing at asampling interval Tm=(Tr×Np) and a sampling frequency fm=1/Tm in thecase where ΔΦ=1/Nc.

Furthermore, by setting Nf to a number that is a power of 2, fastFourier transform processing (FFT) can be applied and the calculationprocessing amount can be greatly reduced. It should be noted that, atsuch time, in the case where Nf>Nc, by performing zero fillingprocessing in which CI(k, Nc(w−1)+q)=0 in regions where q>Nc, likewise,fast Fourier transform processing can be applied and the calculationprocessing amount can be greatly reduced.

It should be noted that, in the aforementioned Doppler frequencyanalysis unit 42, FFT processing may not be performed, and calculationprocessing in which a product sum calculation given by expression (13)is successively performed may be carried out (with respect to CI(k,Nc(w−1)+q+1), which is Nc number of outputs of the integration unit 41obtained at each discrete timepoint k, a coefficient exp[−j2πfsNpqΔφ]corresponding to fs=−Nf+1, . . . , 0, . . . , Nf−1 is generated, andproduct sum calculation processing is successively performed). Here q=0to Nc−1.

Hereinafter, the outputs FT_CI¹(k, fs, w), −, FT_CI^(Na)(k, fs, w) fromthe Doppler frequency analysis unit 42 obtained by the same processingbeing respectively carried out in the first antenna system processingunit to the Na^(th) antenna system processing unit is collectivelydenoted as a correlation vector h(k, fs, w), and is used to describeprocessing in which direction estimation based on phase differencesamong reception antennas is performed with respect to reflected wavesfrom the target.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack & \; \\{{h\left( {k,{fs},w} \right)} = \begin{bmatrix}{{{FT\_}{CI}}^{1}\left( {k,{fs},w} \right)} \\{{{FT\_}{CI}}^{2}\left( {k,{fs},w} \right)} \\\vdots \\{{{FT\_}{CI}}^{Na}\left( {k,{fs},w} \right)}\end{bmatrix}} & (14)\end{matrix}$

It should be noted that, instead of the aforementioned correlationmatrix, a correlation vector may be calculated with one of the pluralityof antenna system processing units serving as a reference phase.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 15} \right\rbrack & \; \\{{h\left( {k,{fs},w} \right)} = {\begin{bmatrix}{{{FT\_}{CI}}^{1}\left( {k,{fs},w} \right)} \\{{{FT\_}{CI}}^{2}\left( {k,{fs},w} \right)} \\\vdots \\{{{FT\_}{CI}}^{Na}\left( {k,{fs},w} \right)}\end{bmatrix}\frac{{FT\_ CI}^{1}\left( {k,{fs},w} \right)^{*}}{{{FT\_ CI}^{1}\left( {k,{fs},w} \right)}}}} & (15)\end{matrix}$

In expression (15), the superscript asterisk (*) indicates a complexconjugate operator, and k=1, . . . , (Nr+Nu)Ns/No.

In the direction estimation unit 43, the correlation vector h(k, fs, w)from the W^(th) number-y Doppler frequency analysis unit 42 output fromthe first antenna system processing unit to the Na^(th) antenna systemprocessing unit is corrected with respect to phase deviation andamplitude deviation among the antenna system processing units using anarray correction value, and a correlation vector h_after_cal(k, fs, w)in which these corrections have been performed is used to performdirection estimation processing based on the phase differences amongreception antennas of arriving reflected waves.

$\begin{matrix}{\mspace{79mu} \left\lbrack {{Equation}\mspace{14mu} 16} \right\rbrack} & \; \\{{h_{{\_ {after}}{\_ {cal}}}\left( {k,{fs},w} \right)} = {\begin{bmatrix}{h\_ cal}_{\lbrack 1\rbrack} & 0 & \ldots & 0 \\0 & {h\_ cal}_{\lbrack 2\rbrack} & \ddots & \ldots \\\vdots & \ddots & \ddots & 0 \\0 & \ldots & 0 & {h\_ cal}_{\lbrack{Na}\rbrack}\end{bmatrix}{h_{y}\left( {k,{fs},w} \right)}}} & (16)\end{matrix}$

In other words, in the direction estimation processing, an azimuthdirection θ indicated in the following expression (17) is made variableusing the correlation vector h_after_cal(k, fs, w) in which phasedeviation and amplitude deviation have been corrected, with respect toeach discrete timepoint k and each Doppler frequency fsΔΦ, or discretetimepoints k and Doppler frequencies fsΔΦ with which the norm ofh_after_cal(k, fs, w) or the square value thereof becomes equal to orgreater than a prescribed value. A direction estimation evaluationfunction value P(θ, k, fs, w) is then calculated, and the azimuthdirection with which the largest value thereof is obtained is taken asan arrival direction estimation value DOA(k, fs, w).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 17} \right\rbrack & \; \\{{{DOA}\left( {k,{fs},w} \right)} = {\arg \; {\max\limits_{\theta_{u}}{P\left( {\theta_{u},k,{fs},w} \right)}}}} & (17)\end{matrix}$

In expression (17), u=1, . . . , NU. It should be noted that arg maxP(x) is an operator with which the value of a domain having the largestfunction value P(x) is taken as an output value.

It should be noted that the evaluation function value P(θ, k, fs, w) isan evaluation function value of various kinds according to the arrivaldirection estimation algorithm. For example, an estimation method thatuses an array antenna disclosed in the literature (‘Direction-of-ArrivalEstimation Using Signal Subspace Modeling’, J. A. Cadzow, Aerospace andElectronic Systems, IEEE Transactions, volume 28, issue 1, publicationyear: 1992, pages 64-79) can be used, and a beam forming method can berepresented by the following expression (18).

[Equation 18]

P(θ_(u) ,k,fs,w)=a(θ_(u))^(H) H _(—after) _(_) _(cal)(k,fs,w)a(θ_(u))  (18)

In expression (18), the superscript H is a Hermitian transpositionoperator. Other than this, it is also possible for techniques such asCapon and MUSIC to be similarly applied.

h_after_cal(k, fs, w) is a correlation matrix, and is given by thefollowing expression (19).

[Equation 19]

H _(—after) _(_) _(cal)(k,fs,w)=h _(—after) _(_) _(cal)(k,fs,w)h_(—after) _(_) _(cal)(k,fs,w)^(H)   (19)

The direction estimation unit 43 then, in addition to the calculatedw^(th) arrival direction estimation value DOA(k, fs, w), uses thediscrete timepoint k, the Doppler frequencies fsΔΦ, and the evaluationfunction value P(DOA(k, fs, w), k, fs, w) of that time as radarpositioning results.

Here, a direction vector a(θ_(u)) is an Na^(th) order column vector inwhich a complex response of an array antenna in the case where radarreflected waves have arrived from an θ_(u) direction are taken aselements. The array antenna complex response a(θ_(u)) represents a phasedifference that is geometric-optically calculated at element intervalsamong antennas. For example, in the case where the element intervals ofthe array antenna are arranged at equal intervals d on a straight line(see FIG. 7), a direction vector can be given by the followingexpression (20).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 20} \right\rbrack & \; \\{{a\left( \theta_{u} \right)} = \begin{bmatrix}1 \\{\exp \left\{ {j\; 2\; \pi \; d\; \sin \; {\theta_{u}/\lambda}} \right\}} \\\vdots \\{\exp \left\{ {j\; 2\; {\pi \left( {N_{a} - 1} \right)}d\; \sin \; {\theta_{u}/\lambda}} \right\}}\end{bmatrix}} & (20)\end{matrix}$

In expression (20), θ_(u) is obtained by causing the azimuth range inwhich arrival direction estimation is performed to change by prescribedazimuth intervals β, and is set as follows, for example. Also, θ_(u)=θmin+uβ, u=0, . . . , NU, and NU=floor[(θmax−θmin)/β]+1. Here, floor(x)is a function that outputs the largest integer value that does notexceed a real number x.

It should be noted that timepoint information may be converted intodistance information and output. The following expression (21) is usedwhen the discrete timepoint k is converted into distance informationR(k).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 21} \right\rbrack & \; \\{{R(k)} = {k\frac{{TwC}\; 0}{2\; L}}} & (21)\end{matrix}$

In expression (21), Tw represents a code transmission segment, Lrepresents a pulse code length, and C0 represents light speed.

Furthermore, Doppler frequency information may be converted into arelative speed component and output. The following expression (22) isused when the Doppler frequency fsΔΦ is converted into a relative speedcomponent vd(fs).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 22} \right\rbrack & \; \\{{{vd}({fs})} = {\frac{\lambda}{2}{fs}\; \Delta \; \theta}} & (22)\end{matrix}$

In expression (22), λ is the wavelength of a carrier frequency of an RFsignal output from the transmission RF unit 25.

Next, a calculation simulation of the aforementioned interferencedetection unit 38 will be described.

The radar device 10 uses a coefficient sequence {FS₁, FS₂, FS₃, FS₄}={1,j, −1, −j} in the frequency component extraction unit 51 to detect afrequency component of a 250-MHz interference signal depicted in FIG. 8,for example, in the radar signal band (500 MHz) of the radar device,with the 1-GSps (giga sample per second) A/D conversion units 37 a and37 b.

Furthermore, another radar device that uses FMCW uses the same carrierfrequency as the radar device 10 to, as depicted in FIG. 9, perform1-GHz frequency sweeping every 10 μs, and cause interference to theradar device 10.

The result of the case where an interference wave level is approximatelythe same as the noise level of the radar device 10 is depicted in FIG.10. From FIG. 10, it is apparent that the output level according to theinterference detection unit 38 increases at a reception timing at whichthe sweeping frequency of the other radar device that uses FMCW becomes250 MHz. Detection sensitivity can also be additionally improved byincreasing the number of times that the coefficient sequence with whichthe frequency component of the interference signal is detected isrepeated.

In this way, according to embodiment 1, an interference measurementsegment is provided in which a radar transmission signal is nottransmitted in the radar transmission unit 20, and, in the radarreception unit 30, detection of a specific frequency component withinthe passband of the radar device 10 is performed in the interferencemeasurement segment for one or more interference signal components to bedetected. In the case where another radar device transmits an FMCW waveas an interference wave, the FMCW wave is frequency-modulated andtherefore has the property that a transmitted frequency componentchanges, and therefore, by detecting the specific frequency componentwithin the passband of the radar device in the interference measurementsegment, it becomes possible to detect interference from the other radardevice.

Furthermore, detection sensitivity can be increased by increasing thenumber of times that a coefficient sequence for extracting the specificfrequency component of the interference detection unit 38 is repeated.Furthermore, the interference detection unit 38 can extract the specificfrequency component by way of a simple circuit configuration withoutusing frequency analysis processing represented by fast Fouriertransform processing, and interference detection can be realized.

Embodiment 2

In embodiment 2 of the present disclosure, the relation betweenfrequency sweeping and radar transmission intervals will be described.

With regard to frequency sweeping periods of another radar device thatuses FMCW, there is a type in which frequency sweeping is performed atcomparatively fast intervals such as of the order of several tens ofmicroseconds (fast frequency modulation type), and a type in whichfrequency sweeping is performed at comparatively slow periods such as ofthe order of milliseconds or the order of several tens of milliseconds.

In the case where the interference measurement segment of the radardevice 10 is longer than the frequency sweeping period of the otherradar device that uses FMCW, and a frequency component included withinthe signal band of the radar device 10 is included within the frequencyrange in which the other radar device performs frequency sweeping,detection becomes possible in one interference measurement segment.

However, even in the case where the interference measurement segment ofthe radar device 10 is shorter than the frequency sweeping period of theother radar device that uses FMCW, and a frequency component includedwithin the signal band of the radar device 10 is included within thefrequency range in which the other radar device performs frequencysweeping, the other radar device sometimes sweeps a frequency componentthat is included within the signal band of the radar device 10 in adistance measurement segment, and there is a possibility of interferencesignal detection failing in the interference measurement segment.

With respect to the aforementioned interference signal detectionfailure, the probability of detecting an interference wave can beincreased by using, in the interference detection unit 38, aconfiguration that detects a plurality of specific frequency componentsincluded within a signal band.

FIG. 11 depicts a configuration for the interference detection unit 38with which two frequency components are detected as specific frequencycomponents. Positive/negative frequency components may be used as afirst frequency component and a second frequency component. For example,as a result of a first frequency component extraction unit 51 using{FS₁, FS₂, FS₃, FS₄}={1, j, −1, −j}, and a second frequency componentextraction unit 71 using {FS₁, FS₂, FS₃, FS₄}={1, −j, −1, j}, specificpositive/negative frequency components ±Ns/(2Tp) can be extracted.

In the case where the interference measurement segment of the radardevice 10 is shorter than the frequency sweeping period of the otherradar device that uses FMCW by approximately 1/D (D being an arbitrarynumber), the probability of detecting an interference wave can beincreased by providing an interference detection unit 38 that detectsapproximately D number of frequency components in substantially equalfrequency intervals within the signal band of the radar device 10.

Embodiment 3

FIG. 12 is a block diagram depicting the configuration of a radar device80 according to embodiment 3 of the present disclosure. FIG. 12 isdifferent from FIG. 1 in that the interference countermeasure controlunit 13 has been removed, the correlation calculation unit 40 has beenchanged to a correlation calculation unit 81, the integration unit 41has been changed to an integration unit 82, the direction estimationunit 43 has been changed to a direction estimation unit 85, and a secondintegration unit 83 and a respective-angle interference componentdetection unit 84 have been added.

The correlation calculation unit 81 performs a correlation calculationin the same way as the correlation calculation unit 40 in interferencemeasurement segments in addition to distance measurement segments. Theintegration unit 82 also performs addition processing in the same way asthe integration unit 41 in interference measurement segments in additionto distance measurement segments.

The second integration unit 83 performs coherent integration withrespect to with floor(N_(IM)/Np) number of outputs from the integrationunit 82 obtained at each discrete timepoint k, with the timings of thediscrete timepoints k being aligned. Here, floor(x) is a function thatoutputs the largest integer that is equal to or less than a real numberx. The second integration unit 83 outputs a coherent integrated resultCCI(k) to the respective-angle interference component detection unit 84.

The respective-angle interference component detection unit 84 usescollected outputs CCI(k) from the second integration unit 83 obtained bythe same processing being respectively carried out in the first antennasystem processing unit to the Na^(th) antenna system processing unit ascorrelation vectors given in the following expressions (23) and (24) toperform direction estimation based on phase differences betweenreception antennas with respect to reflected waves from a target, andcalculates an interference component for each beam angle (hereinafterreferred to as a “respective-angle interference component”) PI(θ_(u)).In the direction estimation processing, calculation processing that usesthe described beam forming method is performed in the directionestimation unit 85.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 23} \right\rbrack & \; \\{h_{I} = {\frac{1}{\bullet \; {{Nr}\bullet}\; {{Nu}\bullet}\; {{Ns}/{No}}}{\sum\limits_{k = 1}^{\bullet \; {{Nr}\bullet}\; {{Nu}\bullet}\; {{Ns}/{No}}}\; {\begin{bmatrix}{{CCI}^{1}(k)} \\{{CCI}^{1}(k)} \\\vdots \\{{CCI}^{Na}(k)}\end{bmatrix}\frac{{{CCI}^{1}(k)}^{*}}{{{CCI}^{1}(k)}}}}}} & (23) \\\left\lbrack {{Equation}\mspace{14mu} 24} \right\rbrack & \; \\{{{PI}\left( \theta_{u} \right)} = {{{{a\left( \theta_{u} \right)}^{H}\begin{bmatrix}{h\_ {cal}}_{\lbrack 1\rbrack} & 0 & \ldots & 0 \\0 & {h\_ {cal}}_{\lbrack 2\rbrack} & \ddots & \ldots \\\vdots & \ddots & \ddots & 0 \\0 & \ldots & 0 & {h\_ {cal}}_{\lbrack{Na}\rbrack}\end{bmatrix}}h_{I}}}^{2}} & (24)\end{matrix}$

In the case where FMCW waves are received as interference from the otherradar device, one or more interference signal components are detected ina substantially uniform manner regardless of the discrete timepoints k,and therefore detection sensitivity for one or more interference signalcomponents can be increased by, in expressions (23) and (24), additionprocessing being performed at the discrete timepoints k (in other words,the distance direction). For example, by performing addition processingfor N number of samples with respect to the discrete timepoints k (inother words, the distance direction), a 5 log₁₀(N) [dB] SNR improvementis achieved. For example, by performing addition processing with respectto 512 samples, an SNR improvement of approximately 13 dB can beachieved.

In the case where it is determined that the output from the interferencedetermination unit 39 in an interference measurement segment includes aninterference component, the respective-angle interference componentdetection unit 84 outputs respective-angle interference componentsPI(θ_(u)) to the direction estimation unit 85. On the other hand, in thecase where it is determined that the output from the interferencedetermination unit 39 does not include an interference component, therespective-angle interference component detection unit 84 outputs therespective-angle interference components PI(θ_(u)) all as zero to thedirection estimation unit 85.

The direction estimation unit 85, in a distance measurement segment,sets a determination threshold value for each angle on the basis of therespective-angle interference components PI(θ_(u)) detected in theinterference measurement segment, with respect to the calculated w^(th)arrival direction estimation value DOA(k, fs, w), the discrete timepointk thereof, the Doppler frequency fsΔΦ, and the evaluation function valueP(DOA(k, fs, w), k, fs, w). In the case where the calculated w^(th)arrival direction estimation value DOA(k, fs, w) is greater thanαPI(θ_(u)), the direction estimation unit 85 outputs the calculatedw^(th) arrival direction estimation value DOA(k, fs, w) as the signal ofthe target detected by the radar device 80. It should be noted that α isa prescribed coefficient value.

According to the aforementioned processing, in an interferencemeasurement segment, an interference component for each angle can bedetected, and a detection determination threshold value can be variablyset for each angle on the basis of the interference power for eachangle. Thus, the probability of an interference component beingerroneously detected as a signal of the target detected by the radardevice 80 can be reduced. Furthermore, in an interference measurementsegment, by performing correlation calculation processing and coherentaddition processing as in a distance measurement segment, a detectiondetermination threshold value can be variably set for each angle inaccordance with the interference state that actually occurs in thedistance measurement segment.

Modified Example 1

The present disclosure is not limited to the configuration of the radardevice of the aforementioned embodiment 2, and may have theconfiguration depicted in FIG. 13. FIG. 13 is a drawing in which theinterference detection unit 38 and the interference determination unit39 have been removed from FIG. 12.

Modified Example 2

The radar transmission signal generation unit 21 is not limited to theconfiguration depicted in FIG. 1, and may have the configurationdepicted in FIG. 14. The radar transmission signal generation unit 21 ofFIG. 14 is provided with a code storage unit 91 and a D/A conversionunit 92. The code storage unit 91 stores code sequences in advance, andsequentially and cyclically reads out the stored code sequences andoutputs to the D/A conversion unit 92.

The D/A conversion unit 92 converts a digital signal output from thecode storage unit 91 into an analog baseband signal and outputs to thetransmission RF unit 25.

The following are included as various aspects of the embodimentsaccording to the present disclosure.

A radar device according to a first disclosure is provided with: areceiver which, in operation, receives one or more radar transmissionsignals transmitted from another radar device, in an interferencemeasurement segment in which transmission of one or more radartransmission signals from the radar device is stopped; an A/D conversioncircuitry which, in operation, converts the one or more radartransmission signals from the other radar device received by thereceiver from one or more analog signals into one or more digitalsignals; and an interference detection circuitry which, in operation,performs a correlation calculation between each of the one or morediscrete samples that is the one or more digital signals and aprescribed coefficient sequence to detect one or more prescribedfrequency components included in the one or more digital signal, as oneor more interference signal components.

The radar device according to a second disclosure is the radar device ofthe first disclosure, in which the interference detection circuitryperforms the correlation calculation using a coefficient sequence inwhich the prescribed coefficient sequence is repeated.

The radar device according to a third disclosure is the radar device ofthe first disclosure, further provided with a transmitter which, inoperation, stops the transmission of the one or more radar transmissionsignals in the interference measurement segment, and transmits the oneor more radar transmission signals in a distance measurement segment inwhich the distance from the radar device to a target is measured.

The radar device according to a fourth disclosure is the radar device ofthe third disclosure, further provided with a transmission controlcircuitry which, in operation, periodically switches between theinterference measurement segment and the distance measurement segment.

The radar device according to a fifth disclosure is the radar device ofthe first disclosure, further provided with interference determinationcircuitry that compares the detected each of the one or moreinterference signal components with a prescribed determination level inthe interference measurement segment, determines that one or moreinterference components are not present when each of the one or moreinterference signal components is equal to or less than thedetermination level, and determines that the one or more interferencecomponents are present when any of the one or more interference signalcomponents exceed the determination level.

The radar device according to a sixth disclosure is the radar device ofthe fifth disclosure, further provided with an interferencecountermeasure control circuitry which, in operation, based on theinterference determination result detected in the interferencemeasurement segment, performs interference countermeasure control in thesubsequent distance measurement segment.

The radar device according to a seventh disclosure is the radar deviceof the sixth disclosure, in which the interference countermeasurecontrol circuitry changes a carrier frequency of the radar device.

The radar device according to an eighth disclosure is the radar deviceof the sixth disclosure, in which the interference countermeasurecontrol circuitry changes the directivity of an antenna of the radardevice for a prescribed time interval.

The radar device according to a ninth disclosure is the radar device ofthe sixth disclosure, in which the interference countermeasure controlcircuitry increases a code length of the one or more radar transmissionsignals in the distance measurement segment for a prescribed timeinterval.

The radar device according to a tenth disclosure is the radar device ofthe first disclosure, in which the prescribed coefficient sequenceincludes a coefficient sequence {1, −j, −1, j} (in which j is animaginary unit).

Heretofore, various embodiments have been described with reference tothe drawings, but it goes without saying that the present disclosure isnot limited to these examples. It is obvious that a person skilled inthe art could conceive of various altered examples or modified exampleswithin the categories described in the claims, and naturally it is to beunderstood that these also belong to the technical scope of the presentdisclosure. Furthermore, the constituent elements in the aforementionedembodiments may be arbitrarily combined without deviating from thepurpose of the disclosure.

In the aforementioned embodiments, the present disclosure has beendescribed with examples in which hardware is used to configure thepresent disclosure; however, it is also possible for the presentdisclosure to be realized also by using software in cooperation withhardware.

Furthermore, each function block used in the description of each of theaforementioned embodiments is typically realized as an LSI, which is anintegrated circuit having an input terminal and an output terminal.These may be implemented separately as single chips or may beimplemented as a single chip in such a way as to include some or all ofthe functional blocks. An LSI has been mentioned here, but a functionblock may be referred to as an IC, a system LSI, a super LSI, or anultra LSI depending on the difference in the degree of integration.

Furthermore, the circuit integration technique is not limited to an LSI,and a function block may be realized using a dedicated circuit or ageneral-purpose processor. After an LSI is manufactured, afield-programmable gate array (FPGA) that can be programmed, or areconfigurable processor with which the connections and settings ofcircuit cells within an LSI can be reconfigured, may be used.

In addition, if circuit integration technology that replaces LSI appearsas a result of another technology that is an advancement insemiconductor technology or is derived therefrom, naturally, the othertechnology may be used to carry out the integration of function blocks.The application and so forth of biotechnology is also a possibility.

A radar device according to the present disclosure can be applied to amoving body including a vehicle.

What is claimed is:
 1. A radar device, comprising: a receiver which, inoperation, receives one or more radar transmission signals transmittedfrom another radar device, in an interference measurement segment inwhich transmission of one or more radar transmission signals from theradar device is stopped; A/D conversion circuitry which, in operation,converts the one or more radar transmission signals from the other radardevice received by the receiver from one or more analog signals into oneor more digital signals; and interference detection circuitry which, inoperation, performs a correlation calculation between each of the one ormore discrete samples that is the one or more digital signals and aprescribed coefficient sequence to detect one or more prescribedfrequency components included in the one or more digital signals, as oneor more interference signal components.
 2. The radar device according toclaim 1, wherein the interference detection circuitry performs thecorrelation calculation using a coefficient sequence in which theprescribed coefficient sequence is repeated.
 3. The radar deviceaccording to claim 1, further comprising: a transmitter which, inoperation, stops the transmission of the one or more radar transmissionsignals in the interference measurement segment, and transmits the oneor more radar transmission signals in a distance measurement segment inwhich a distance from the radar device to a target is measured.
 4. Theradar device according to claim 3, further comprising: transmissioncontrol circuitry which, in operation, periodically switches between theinterference measurement segment and the distance measurement segment.5. The radar device according to claim 1, further comprising:interference determination circuitry which, in operation, compares eachof the detected one or more interference signal components with aprescribed determination level in the interference measurement segment,determines that one or more interference components are not present wheneach of the one or more interference signal components is equal to orless than the determination level, and determines that the one or moreinterference components are present when any of the one or moreinterference signal components exceeds the determination level.
 6. Theradar device according to claim 5, further comprising: interferencecountermeasure control circuitry which, in operation, based on theinterference determination result detected in the interferencemeasurement segment, performs interference countermeasure control in thesubsequent distance measurement segment.
 7. The radar device accordingto claim 6, wherein the interference countermeasure control circuitrychanges a carrier frequency of the radar device.
 8. The radar deviceaccording to claim 6, wherein the interference countermeasure controlcircuitry changes directivity of an antenna of the radar device for aprescribed time interval.
 9. The radar device according to claim 6,wherein the interference countermeasure control circuitry increases acode length of the one or more radar transmission signals in thedistance measurement segment for a prescribed time interval.
 10. Theradar device according to claim 1, wherein the prescribed coefficientsequence includes a coefficient sequence {1, −j, −1, j} (in which j isan imaginary unit).